This invention relates to conductivity control of capacitive input power switching devices.
Power switching devices having a capacitive gate control input (e.g., MOSFET, insulated gate bipolar transistor (IGBT), MOS-controlled thyristor (MCT)) are used in a multitude of electronic switching applications (e.g., on/off load controls, switching amplifiers, motor drivers, DC--DC converters, cycloconverters). Such devices are turned on by charging the gate capacitance to some appropriate, relatively low, voltage value (e.g., 8 or 15 volts) and are turned off by discharging the gate capacitance (e.g., to a voltage near zero volts). Information as to when the power switching device is to be on and off is typically generated by low-level electronic control circuitry. This information is delivered to a gate driver circuit which is designed to rapidly charge and discharge the gate capacitance of the power switching device as a means of turning the device on and off. The capacitive characteristic of the gate terminal implies that turn-on and turn-off speed of the power switching device will be dependent, in large part, upon the ability of the gate driver to source and sink current into and out of the gate capacitance.
One conventional way to turn such a power switching device on and off is by means of a pair of gate switches (FIG. 1). A first gate switch 12 is connected in series with a source of gate bias voltage 16 and the gate terminals of the power switching device 10; a second gate switch 14 is connected across the gate terminals. Turning the first switch on (by means of the control circuitry 18), with the second switch off, charges the gate capacitance 20 to a voltage essentially equal to that of the bias voltage source; turning the second switch on (with the first switch off) discharges the gate capacitance to essentially zero volts. In a conventional gate driver circuit 22 of this kind the total energy dissipated in the gate driver circuit during each switching cycle (i.e., turn-on/turn-off cycle) will be equal to twice the peak energy stored in the gate capacitance 20 during the cycle: half of this energy is lost in the first gate switch 12 during charging, the other half is lost in the second gate switch 14 during discharge. As the switching frequency (i.e., the rate at which switching cycles occur) increases, the amount of power dissipated in the gate driver will increase proportionately. This can be particularly troublesome in high-frequency power switching applications.
Steigerwald (U.S. Pat. No. 4,967,109, Oct. 30, 1990) shows how to improve upon the performance of a conventional gate driver by placing a diode and inductor in series with the first gate switch (FIG. 2). When the first gate switch 22 is turned on (with the second gate switch 24 off) the inductor 32 resonates with the gate capacitance 30 and the gate voltage "rings up" to a voltage equal to twice the bias source 26 voltage. The diode 34 prevents reverse energy flow back from the gate capacitance to the inductor. The gate capacitance is discharged by turning the second gate switch 24 on (with the first gate switch off). There are several benefits to this quasi-resonant gate driver 36 approach: (a) by using a quasi-resonant energy transfer mechanism, the loss in the first gate switch is essentially eliminated, yet a fast transition in the rise of gate voltage can be achieved by appropriate selection of the inductor value (i.e., by setting the equivalent characteristic time constant of the LC circuit formed by the gate capacitance 30 and the inductor 32 to a suitably small value) and, (b) the value of the bias voltage source needed to achieve some desired value of gate turn-on voltage can be reduced by one half. In U.S. Pat. No. 5,010,261, Apr. 23, 1991, Steigerwald discloses a more complex gate drive circuit which provides quasi-resonant transitions at both turn-on and turn-off, further reducing losses in the driver. A related approach, using a single switch across the gate of the power switching device and an inductor and diode in series with the gate and a bias voltage source, is described by Tabisz, Gradzki, et al., in "Zero-Voltage Switched Quasi-Resonant Buck and Flyback Converters--Experimental Results at 10 MHz," IEEE Transactions on Power Electronics, April 1989, p. 194. In the Tabisz circuit quasi-resonant ring-up of the gate voltage to a value more than twice the value of the bias voltage source is achieved by use of a "flyback" approach in a constant frequency power converter.
The conventional and quasi-resonant gate driver circuits 22, 36 described above are examples of circuits in which the gate driver, the control circuitry and the gate terminals of the power switching device are all referenced to a common signal return. There are many applications, however, in which the gate terminals of the power switching device are referenced to a circuit node which "floats" at a voltage value different from that of the control circuitry signal return. For example, in FIG. 3, the gate voltage, Vg, of the "high side" power switching device 40 is referenced to a circuit node 42 which might vary by hundreds of volts as the power switching device turns on and off. One way to drive the gate of a floating power switching device is to use a non-isolated active level-shifting gate driver circuit. One such circuit, manufactured as a monolithic integrated circuit, part No. IR2125, by International Rectifier, California, U.S., is described in Preliminary Data Sheet No. PD-6.017, "High Voltage Current Limiting MOS Gate Driver". Other active level-shifting circuits are described in International Rectifier Application Note AN-978A, 1990. Such circuits have a number of drawbacks: (a) they are lossier and more complex than conventional non-floating drivers; (b) the active level-shifting devices used in such circuits (e.g., semiconductor devices) must withstand the voltage difference between the floating node and the control circuitry signal return, and these devices may fail catastrophically if this voltage is allowed, even momentarily, to exceed the device's breakdown voltage rating; (c) the lack of galvanic isolation between the control circuitry and the floating gate increases circuit sensitivity to the effects of noise and rate-of-change of the floating voltage, and (d) they require a floating source of gate bias voltage (e.g., a floating bias supply, a "bootstrap" circuit, a charge pump).
Another way of implementing a floating gate driver is to incorporate an isolation transformer in the gate driver circuit. One such gate driver circuit is disclosed in Barzegar, U.S. Pat. No. 4,748,351, "Power MOSFET Gate Driver Circuit", May 31, 1988 (and shown schematically in FIG. 4). In the Figure, the transformer isolated gate driver 50 includes a conventional pulse isolation transformer 52 with a primary winding 54 referenced to control circuit return and a pair of secondary windings 56, 58 referenced to the floating gate terminal node 60. A differential driver 62 connected to the primary winding allows changing the polarity of the voltage pulses delivered by the secondary windings. One secondary winding 56 is connected in series with a first diode 64 and the gate terminals of the power switching device 66; the other secondary winding 58 is connected in series with a second diode 68 and the gate control terminals of a gate switch 70 which is wired across the gate terminals of the power switching device 66. When the primary is driven by a pulse of a first polarity, the gate capacitance 72 of the power switching device is charged to a voltage determined by the voltage output of the differential driver and the turns ratio of the isolation transformer 52; when the primary is driven by a pulse of the opposite polarity, the gate switch 70 turns on and the gate capacitance of the power switching device is discharged. The transformer isolated gate driver 50 of FIG. 4 improves upon active level-shifting in terms of reduced complexity, improved breakdown voltage capability, improved noise immunity and elimination of the need for a floating bias source. Other transformer isolated driver configurations are illustrated in the previously referenced International Rectifier application note. From a loss viewpoint, however, none of the referenced transformer-isolated circuits are an improvement over a conventional gate driver circuit.
Galvanically isolated gate driver circuits using optical couplers are also described in the prior art. However, such circuits generally require a floating source of bias voltage and exhibit propagation delays which are typically longer than those found in electrically coupled circuits.
In U.S. Pat. No. 4,415,959, Vinciarelli teaches how energy may be losslessly transferred from an input voltage source to a capacitor via a controlled amount of leakage inductance in an isolation transformer. In Vinciarelli's forward power converter circuit a switch couples a voltage source to the primary of a transformer having a controlled amount of leakage inductance. The secondary winding of the transformer is connected in series with a capacitor and a diode. The switch is closed at zero current; current rises and falls essentially sinusoidally in the transformer windings as a quantum of energy is passed from the input source to the capacitor via the transformer leakage inductance; and the switch is opened at zero current. During this energy transfer phase, and assuming ideal components, the capacitor will charge to a voltage essentially equal to twice the reflected source voltage (the reflected source voltage being equal to the input source voltage multiplied by the coupling coefficient of the tranformer and divided by the transformer primary to secondary turns ratio). The diode in series with the secondary prevents any of the energy stored in the capacitor from returning back to the source.